Method and system for optical impairment mitigation for high-speed optical communication systems

ABSTRACT

A method and system for simultaneous mitigation of optical impairment from both equalizer-phase noise interaction (EPNI) and fiber nonlinear effects (FNE) is disclosed. In one embodiment, the method is directed to simultaneous mitigation of optical impairment from both equalizer-phase noise interaction (EPNI) and fiber nonlinear effects (FNE) using a fast-adaptive multi-tap digital filter.

This application is a continuation of U.S. patent application Ser. No.14/017,433, filed Sep. 4, 2013, the disclosure of which is incorporatedby reference herein in its entirety.

BACKGROUND

The popularity of multimedia communications services over packet datanetworks, such as the Internet, continues to grow. Consequently, thedemand for higher capacity in core data transport networks continues togrow. For service providers, core data transport networks are typicallyoptical networks based on fiber optic technology. To meet theever-growing capacity demand, 100 G/s per channel data rate operating at2 bit/s/Hz spectral efficiency (SE) by using quadrature phase shiftkeying in the pulse-density (PDM-QPSK) modulation and digital signalprocessing (DSP) enabled coherent detection has been commerciallydeployed in the existing core networks. Also, research for the nextgeneration of transport systems operating at even higher data rate(likely at 400 Gb/s or above) and higher SE by using more spectrallyefficient high-order modulation formats (such as the well knownquadrature amplitude modulation (QAM)) is underway. However, recentexperimental results have revealed that it is extremely challenging toachieve long-reach transmission using high-order modulation formatsbecause they are more vulnerable to various transmission impairmentssuch as fiber nonlinear effects, laser phase noise, and amplifier noise.

To address the challenge caused by fiber nonlinear effects, severaldigital nonlinear compensation methods have been proposed, including thedigital backward-propagation (DBP) based methods and a Volterra-basednonlinear equalization method. However, the implementation complexity ofthese digital methods is prohibitively high, making it almost impossiblefor them to be realized for any practical high-speed transmissionsystems. Several mid-link phase-conjugation based methods have also beenproposed. However, these methods generally work well only forpoint-to-point submarine systems or specially designed super-channelsystems. For typical terrestrial optical networks where reconfigurableoptical add/drop multiplexers (ROADMs) are used to route opticalwavelengths and different wavelength channels usually end up atdifferent locations, there is still no feasible solution to mitigatefiber nonlinear effects for high-speed coherent optical transmissionsystems.

To reduce the impact of laser phase noise on system performance, severalsingle-tap phase rotation filter (with fast adaptive rate) based phaserecovery methods have been proposed. However, these methods only workwell for systems without using long-memory equalizer at the receiver,i.e. the short-reach system or long-reach system using inline opticaldispersion compensation. Because the use of inline optical dispersioncompensation not only increases the complexity of the inline opticalamplifier design, but also significantly reduces fiber nonlineartolerance, purely electrical/digital dispersion compensation is usuallyrequired in a high-speed coherent optical transmission system. For sucha communication system, a linear digital filter/equalizer with very longmemory length has to be introduced at the receiver to compensate for theaccumulated dispersion from the transmission fiber.

BRIEF SUMMARY

It has been determined that the use of long-memory filter/equalizer atthe receiver (or at the transmitter for pre-compensation) will not onlyenhance the received phase noise, but will also convert the phase noiseinto amplitude noise, causing additional signal distortion which cannotbe mitigated using the traditional carrier phase recovery method.Moreover, it has been determined that the penalty from such anequalizer-phase noise interaction increases with the data rate and canpose a problem for future 400-Gb/s systems operating at a very high baudrate. To address this problem, a hardware-based laser phase noisecompensation method has recently been proposed. However, this method isvery complex and costly because it requires an additional set ofcoherent receiver to measure the laser phase noise.

The present disclosure is generally directed to a DSP-based solution toreduce transmission impairments such as fiber nonlinear effects, laserphase noise, and amplifier noise for long-reach transmission of signalsusing high-order modulation formats. In one embodiment, a method andsystem is directed to a single fast-adaptive multi-tap digital filterfor simultaneous mitigation of optical impairment from bothequalizer-phase noise interaction (EPNI) and fiber nonlinear effects(FNE). In another embodiment, the method and system is directed to theuse of single fast-adaptive multi-tap digital filter to perform thecommon phase recovery function in addition to simultaneous mitigation ofoptical impairment from both EPNI and FNE.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a DSP-enabled coherent optical communication system,according to one embodiment;

FIG. 2 illustrates a system for optical impairment mitigation forhigh-speed optical communication systems, according to one embodiment;

FIG. 3 illustrates a flow-chart showing a method for optical impairmentmitigation for high-speed optical communication systems, according toone embodiment;

FIG. 4 illustrates an exemplary flow-chart showing a block-by-blockbased feed-forward adaption solution for fast adaption of thefast-adaptive multi-tap digital filter, according to one embodiment;

FIG. 5 illustrates an exemplary Bit-Error-Rate performance of aWavelength-Division Demultiplexing system with and without using of thefast-adaptive multi-tap digital filter, according to one embodiment;

FIG. 6 shows exemplary simulated constellation diagrams for theWavelength-Division Demultiplexing system with and without the use ofthe fast-adaptive multi-tap digital filter; and

FIG. 7 illustrates a high-level block diagram of an exemplary computerthat may be used for implementing a DSP-enabled coherent opticalcommunication system and a method for optical impairment mitigationusing the fast-adaptive multi-tap digital filter.

DETAILED DESCRIPTION

The present disclosure is directed to a method and system forsimultaneous mitigation of optical impairment from both equalizer-phasenoise interaction (EPNI) and fiber nonlinear effects (FNE). In oneembodiment, the method is directed to simultaneous mitigation of opticalimpairment from both equalizer-phase noise interaction (EPNI) and fibernonlinear effects (FNE) using a fast-adaptive multi-tap digital filter.

FIG. 1 illustrates a DSP-enabled coherent optical communication systemaccording to one embodiment. Specifically, FIG. 1 illustrates aDSP-enabled coherent optical communication system where a transmitterlaser 102 with carrier frequency w_(s) and phase noise θs(t) ismodulated by an optical modulator 104. The modulated optical signal isthen transmitted through an optical-amplified transmission link 106 tocoherent mixer 110 where the optical signal is coherently mixed with acontinuous wave (CW) light emitted from a local oscillator (LO) 108 withcarrier frequency w_(L) and phase noise denoted as θL(t). After mixingin coherent mixer 110, the in-phase and quadrature components at twoorthogonal polarizations (nominally denoted as X-pol and Y-pol) of thereceived signal are converted, using an optical-to-electrical converter112, into analog electrical signals, which are then digitized byanalog-to-digital converter 114. The digitized signals 116 are then sentto a DSP unit 118 for signal recovery and demodulation.

The process of signal recovery and demodulation using DSP unit 118consists of five steps. First, the received digitized signal 116 at eachpolarization is sent to a long-memory ‘static’ linear filter/equalizer120 for fiber chromatic dispersion (CD) compensation. In one embodiment,as shown in FIG. 1, the received digitized signal 116 could be presentedas a plurality of signals, each signal having its own polarization. Inthe case of the plurality of signals, the DSP receiver may include aplurality of the long-memory ‘static’ linear filter/equalizers 120, eachof the long-memory ‘static’ linear filter/equalizers 120 configured toreceive and process the signal having its own polarization. In anotherembodiment, the received digitized signal 116 could be presented as asingle signal having separate channels for each polarization (notshown). In the case of the single signal, a DSP receiver may include along-memory ‘static’ linear filter/equalizer 120 may be a singlelong-memory ‘static’ linear filter/equalizer 120 configured to receiveand process a single signal having separate channels for eachpolarization. It is to be understood that in the case of the singlesignal, channels occupy different frequency bands.

The CD-compensated signals 122 then pass through a butterfly-configured2×2 adaptive equalizer 124 for polarization recovery and residualdispersion compensation. It is to be understood that the CD-compensatedsignals 122 may be presented as either a plurality of signals, as shownin FIG. 1, or as a single signal having a plurality of channels (notshown). Because the rate of polarization change is much slower than thesymbol rate (4 to 6 orders of magnitude slower), the required adaptiverate can be relatively slow, allowing the use of decision-directedadaptive equalization algorithms even with the use of high degree oftime-interleave based parallel processing. Carrier frequency and phaserecovery are performed following the polarization recovery. For carrierfrequency recovery, the carrier frequency offset between the signalsource and the LO, i.e., Δw in FIG. 1, is estimated using a frequencyoffset estimator 126 and then removed using a one-tap phase-rotationfilter 128. For carrier phase recovery, the total phase noise (from thesignal source and the LO) is estimated by using one of severalalgorithms, such as described in the “Nonlinear estimation ofPSK-modulated carrier phase with application to burst digitaltransmission,” A. J. Viterbi and A. M. Viterbi, IEEE Trans. Inf. Theory,vol. IT-29, no. 4, July 1983, “An improved feed-forward carrier recoveryalgorithm for coherent receiver with M-QAM modulation format,” X. Zhou,IEEE Photonics Technol. Lett., Vol. 22, No. 14, pp. 1051-1053, Jul. 15,2010, and “Phase-Noise-Tolerant Two-Stage Carrier Recovery Concept forHigher Order QAM Formats,” T. Pfau and R. No´ e, IEEE J. SELECTED TOPICSIN QUANTUM ELECTRONICS, VOL. 16, NO. 5, pp. 1210-1216, September/October2010, using a phase estimator 130 and then removed using another one-tapphase rotation filter 132. The phase-recovered signal 134 is then sentto the decision-making unit 136 for final signal demodulation.

The DSP functions described above work well for a liner transmissionsystem using extremely narrow line-width lasers. For a practical fibertransmission system using a narrow line width laser operating in therange from 100 kHz to 1 MHz, however, there exist two major problems.First, the fiber system is not a linear system, and the Kerr nonlineareffect will cause additional signal distortion which is not compensatedfor or mitigated by using the conventional coherent receiver design asshown in FIG. 1. Secondly, the long-memory filter used in the receiverfor the CD compensation will not only enhance the LO phase noise, butalso convert the LO phase noise into amplitude noise. The long-memoryCD-compensating filter will not enhance the phase noise from the signalsource because the signal source passes through both the transmissionfiber and the CD-compensating filter. For the case of perfect CDcompensation, the fiber dispersion and the CD-compensating filter willcancel out the impact from each other. On the contrary, the LO onlypasses through the CD-compensating filter so the impact from this filtercannot be canceled out. Such additional signal distortion caused by theinteraction between the equalizer and the LO phase noise cannot becompensated for or mitigated by using the conventional one-tapphase-rotation filter as is shown in FIG. 1.

The signal distortion caused by both EPNI and FNE are correlated overmultiple symbol periods. The symbol period is the time-domain durationfor each data pulse. A time-varying multi-tap linear filtering processcan be used to model both EPNI and FNE (to the first order). In oneembodiment, a fast-adaptive multi-tap digital filter replaces theconventional one-tap phase rotation filter. Such a fast-adaptivemulti-tap filter performs not only the normal phase recovery function,but also helps reduce the penalty due to additional signal distortioncaused by EPNI and FNE.

FIG. 2 illustrates a DSP-enabled coherent optical communication systemfor optical impairment mitigation using the fast-adaptive multi-tapdigital filter according to one embodiment. Transmitter laser 202 withcarrier frequency w_(s) and phase noise θs(t) is modulated by an opticalmodulator 204 of a transmitter 201. The modulated optical signal is thentransmitted through an optically-amplified transmission link 206 tocoherent mixer 210 where the modulated optical signal is then coherentlymixed with a CW light emitted from a local oscillator (LO) 208 withcarrier frequency w_(L) and phase noise denoted as θL(t). After mixingin coherent mixer 210, the in-phase and quadrature components at twoorthogonal polarizations (nominally denoted as X-pol and Y-pol) of thereceived signal are converted into analog electrical signals using anoptical-to-electrical converter 212. The electrical signals are thendigitized using analog-to-digital converter 214. The digitized signals316 are sent to a DSP unit 218 for signal recovery and demodulation.

In the DSP unit 218 the received digitized signal 216 at eachpolarization is sent to a long-memory ‘static’ linear filter/equalizer220 for fiber chromatic dispersion (CD) compensation. The CD-compensatedsignals 222 are then passed through a butterfly-configured 2×2 adaptiveequalizer 224 for polarization recovery and residual dispersioncompensation. As noted above, because the rate of polarization change(<1 MHz) is much slower than the symbol rate (10-100 GHz, 4 to 6 ordersof magnitude slower), the required adaptive rate can be relatively slow,allowing the use of decision-directed adaptive equalization algorithmseven with the use of high degree of time-interleave based parallelprocessing. Carrier frequency and phase recovery are performed followingthe polarization recovery. For carrier frequency recovery, the carrierfrequency offset between the signal source and the LO, (i.e., Δw) isestimated using a frequency offset estimator 226 and then removed fromthe signal outputted from the butterfly-configured 2×2 adaptiveequalizer 224 using a one-tap phase-rotation filter 228. The total phasenoise is estimated by using a fast-adaptive multi-tap digital filter232. Because laser phase noise typically varies 2 to 4 orders ofmagnitude faster than the state of polarization change (tens ofmicroseconds versus tens of nanoseconds), the adaption rate for thefast-adaptive multi-tap digital filter 232 should be much faster thanthe regular polarization equalizer. A high adaptation rate can berealized using feed-forward based adaptation algorithms, such as, ablock-by-block least square (LS) based algorithm, where the receiveddata stream is divided into consecutive blocks, and filter coefficientsof the said multi-tap filter are assumed to be constant over each datablock, which may consist of tens to hundreds of consecutive datasymbols. The changes of the filter coefficients from one block to thefollowing block are estimated using LS based algorithms. To reduce theimpact of imperfect decision accuracy, multiple iterations may beapplied to each data block for filter coefficients update. Thephase-recovered signal 234 is then sent to the decision-making unit 236for final signal demodulation.

FIG. 3 illustrates a method for an optical impairment mitigation using afast-adaptive multi-tap filter in the DSP-enabled coherent opticalcommunication system as shown in FIG. 2. At step 302, a firstcompensated digitized signal is generated using a linear filter.Specifically, the first compensated digitized signal is generated bysending a received digitized signal at each polarization to along-memory ‘static’ linear filter/equalizer for fiber chromaticdispersion (CD) compensation. In one embodiment, the received digitizedsignal is signal 216 of FIG. 2 and the long-memory ‘static’ linearfilter/equalizer for fiber chromatic dispersion (CD) compensation is oneor more 1×1 CD equalizers 220 of FIG. 2.

At step 304, a second compensated signal is generated by conducting apolarization recovery and residual dispersion compensation using anadaptive equalizer. In one embodiment, the polarization recovery andresidual dispersion compensation is performed by passing the firstcompensated digitized signal through a butterfly-configured 2×2 adaptiveequalizer. In one embodiment, step 304 is presented in FIG. 2 as signal222 being transmitted to butterfly-configured 2×2 adaptive equalizer 224for polarization recovery and residual dispersion compensation where thesignal output from the butterfly-configured 2×2 adaptive equalizer 224is the second compensated signal of step 304. It should be noted thatbecause the rate of polarization change is much slower than the symbolrate (4 to 6 orders of magnitude slower), the required adaptive rate canbe relatively slow, allowing the use of decision-directed adaptiveequalization algorithms even with the use of high degree oftime-interleave based parallel processing. In one embodiment, theadaptation algorithm processor 238 of FIG. 2 runs the decision-directedadaptive equalization algorithm for the polarization recovery andresidual dispersion compensation. At step 306, a frequency offset of thesecond compensated signal is estimated. In one embodiment, step 306 isperformed as part of a carrier frequency recovery performed followingthe polarization recovery. In one embodiment, for the carrier frequencyrecovery, the carrier frequency offset Δw between the signal source andthe LO is estimated using frequency offset estimator 226 of FIG. 2.

At step 308, the frequency-recovered signal is generated by removing thefrequency offset of the second compensated signal. In one embodiment,the carrier frequency offset Δw is removed by one or more single-tapfilters 228 of FIG. 2. In one embodiment, the removal of the carrierfrequency offset Δw in step 308 results in generation offrequency-recovered signal 240 of FIG. 2.

At step 310, a phase-recovered signal is generated by performing a phaserecovery of the frequency-recovered signal using a fast-adaptivemulti-tap digital filter. As noted above, the adaption rate for theproposed fast-adaptive multi-tap digital filter should be much fasterthan the regular polarization equalizer because laser phase noisetypically varies 2 to 4 orders of magnitude faster than the state ofpolarization change (tens of microseconds versus tens of nanoseconds).In one embodiment, a fast adaptation rate is achieved by usingfeed-forward based adaptation algorithms run on fast adaptive algorithmprocessor 230 of FIG. 2. In one embodiment, a classic block-by-blockleast square (LS) based algorithm is used as the feed-forward basedadaptation algorithm.

The method for an optical impairment mitigation of FIG. 3 concludes withstep 312 where the phase-recovered signal generated at step 310 isdemodulated.

FIG. 4 illustrates an exemplary block-by-block based feed-forwardadaption solution for fast adaption of the fast-adaptive multi-tapdigital filter 232 of FIG. 2.

At step 402, the frequency-recovered signal, generated at step 308 ofFIG. 3, is divided into blocks having multiple overlap symbolsintroduced between the blocks. At step 404, a carrier phase over eachblock is estimated using common phase estimation algorithms, where thephase noise is assumed to be a constant over a multiple symbol periodand it is also assumed that there is negligible phase noise to amplitudenoise conversion. At step 406, an estimated carrier phase is removedusing the fast-adaptive multi-tap digital filter 232 of FIG. 2.

At step 408, an initial decision is made. A person skilled in the artwould understand that “the decision” means the process in which areceiver determines the value of transmitted symbols in a signal. Forexample, if the transmitter sends “10101” to the receiver by pulseamplitude modulation, due to the noise corruption, the receiver needs tomake a decision as to which pulse symbol has a value of “1” and whichpulse symbol has a value of “0.” In one embodiment, the initial decisionis made based on performing strictly phase recovery over a current datablock. But the initial decision may also be made by applying therecovered phase of a prior data block to the current data block or bydirectly applying the EPNI/FNE filter coefficients acquired from theprior data block to the current data block, where the starting phase orEPNI/FNE coefficients may be obtained using a starting trainingsequence. Since the block length cannot be too large due to the need forrapid adaption, accumulated amplifier noise may degrade the performanceof the fast-adaptive multi-tap digital filter. This drawback may bealleviated by joint optimization of the fast-adaptive multi-tap digitalfilter at both polarizations because the phase noise in X- andY-polarization is usually correlated (since they are typically from thesame source).

At step 410, using a signal initially decided at step 408 as a referencesignal, one or more ‘optimal’ coefficients of the fast-adaptivemulti-tap digital filter are estimated by using for example well-knownLS based algorithms.

At step 412, ‘optimal’ coefficients of the fast-adaptive multi-tapdigital filter are updated with the one or more optimal coefficientsestimated at step 410.

At step 414, the frequency-recovered signal of step 308 of FIG. 3 isequalized using the fast-adaptive multi-tap digital filter havingupdated ‘optimal’ coefficients. At step 416, a decision is made by thedecision-making unit 236 with respect to the frequency-recovered signalequalized at step 414. It is to be understood that, to reduce the impactof imperfect decision accuracy, multiple iterations may be applied toeach data block for filter coefficients update.

It should be noted that the method of FIG. 3 may also be used for thecase where the long-memory CD-compensation filter is placed at thetransmitter (i. e. pre-compensation). For the case where the long-memoryCD-compensation filter is placed at the transmitter, there is no longmemory filter placed at the receiver, so the LO will not be affected bythe impact of long-memory filter. However, the transmitter source willexperience only the fiber dispersion and the impact from fiberdispersion cannot be canceled out. Accordingly, in this case, theinteraction between the fiber dispersion and the signal source phasenoise will be similar to the interaction between the LO and theCD-compensating filter and therefore can also be mitigated by using theproposed method.

It should also be noted that the method of FIG. 3 can be easily extendedto future space division multiplexing (SDM) systems, where not only along-memory filter is required for fiber CD compensation, but amulti-input multi-output (MIMO) equalizer having a substantial lengthmay also be needed (e.g. the use of few-mode or coupled multi-corefibers) for modal dispersion compensation. In this case, the impairmentcaused by the interaction between the long MIMO equalizer and the laserphase noise as well as the impairments caused by inter-mode nonlineareffects may be compensated by the proposed fast-adaptive multi-tapequalization method. It should be noted that the use of multiple spatialmodes can be used to enable the improvement of the EPNI/FNE equalizationperformance by joint optimizing the equalizer coefficients over multiplespatial modes.

The impairment mitigation method described above has been numericallyverified for a 7-channel 50 GHz-spaced 49 Gbaud PDM-16QAM system(operating at 392-Gb/s per channel bit rate, with Nyquist pulse shapingusing a roll off factor 0.01) by using a block-by-block iterative LSalgorithm. The transmission link consists of total 20erbium-doped-fiber-amplified (EDFA) spans, and each span is composed of100 km of large area fiber (dispersion coefficient and fiber loss areassumed to be 21 ps/nm/km and 0.18 dB/km, respectively) and EDFA-onlyamplification (noise figure is assumed to be 5 dB). No inline opticaldispersion compensation is used for this simulation. For simplicity,polarization-mode dispersion (PMD) and polarization-dependent loss isnot considered in this simulation. For the laser sources, we assume thatthe signal source and the LO have identical line width.

FIG. 5 illustrates an exemplary Bit-Error-Rate performance of aWavelength-Division Demultiplexing system with and without using of thefast-adaptive multi-tap digital filter. In an exemplary embodiment, FIG.5 shows the bit error ratio (BER) performance of the middle channel (ch.4) versus the laser line width at the optimal signal launch power 3dBm/channel. Line 502 depicts the results generated using a conventionalcoherent receiver with a sliding-window based two-stage maximumlikelihood phase recovery algorithm, while line 504 depicts the resultgenerated by the proposed EPNI/FNE mitigation method using thefast-adaptive multi-tap digital filter, where a 5-tap Ts-spaced (whereTs denotes the symbol period) 1×1 linear equalizer operating with ablock-by-block adaptive LS algorithm is used at each polarization forsimultaneous phase recovery and additional EPNI/FNE distortionmitigation. In one embodiment, the block length is chosen to be 80symbols (including 5 overlap symbols) and three iterations are appliedfor each data block, where the initial decision for each data block ismade based on the same phase recovery algorithm used for theconventional coherent receiver (i.e. for the blue symbols).

Based on the results illustrated on FIG. 5, a person skilled in the artwill understand that the proposed method effectively mitigates theimpairments caused by EPNI and FNE. For the case that there is no phasenoise, i.e. the laser line width is zero, the proposed method improvesthe Q performance by 0.25 dB by mitigating the impairments caused byFNE. For a laser line width 0.8 MHz (a typical line width for the widelyused DFB laser), the proposed method improves the Q performance by 1.15dB by mitigating the impairments from both EPNI and FNE. A 1.1 dB Qperformance improvement can translate into a transmission reach increaseby approximately 30%. The performance improvement of the proposed methodcan also be clearly seen from the constellation diagrams shown in FIG. 6where constellation diagram 601 illustrates the results generated usinga conventional coherent receiver with a sliding-window based two-stagemaximum likelihood phase recovery algorithm and where constellationdiagram 602 illustrates the results generated by the proposed newEPNI/FNE mitigation method using the fast-adaptive multi-tap digitalfilter.

It is to be understood that the proposed fast-adaptive multi-tap digitalfilter can be implemented as a 1×1 linear filter for each polarization,or each spatial mode for an SDM system, where the filter coefficientsmay be optimized either independently for each polarization/spatial modeor optimized by jointly considering more than one polarizations/spatialmodes. Furthermore, the proposed fast-adaptive multi-tap digital filtermay also be implemented as a butterfly-configured N×N MIMO equalizer forjoint optimization of multiple spatial modes (two orthogonalpolarizations can be assumed as two spatial modes).

FIG. 7 illustrates a high-level block diagram of an exemplary computerthat may be used for implementing a new DSP-enabled coherent opticalcommunication system and a method for optical impairment mitigationusing the fast-adaptive multi-tap digital filter. Computer 700 comprisesa processor 701 operatively coupled to a data storage device 702 and amemory 703. Processor 701 controls the overall operation of computer 700by executing computer program instructions that define such operations.The computer program instructions may be stored in data storage device702, or other computer readable medium, and loaded into memory 703 whenexecution of the computer program instructions is desired. Thus, thesteps of FIGS. 4 and 5 can be defined by the computer programinstructions stored in memory 703 and/or data storage device 702 andcontrolled by processor 701 executing the computer program instructions.For example, the computer program instructions can be implemented ascomputer executable code programmed by one skilled in the art to performan algorithm defined by the method steps of FIGS. 4 and 5. Accordingly,by executing the computer program instructions, the processor 701executes an algorithm defined by the method steps of FIGS. 4 and 5.Computer 700 also includes one or more network interfaces 705 forcommunicating with other devices via a network. Computer 700 alsoincludes one or more input/output devices 704 that enable userinteraction with computer 700 (e.g., display, keyboard, mouse, speakers,buttons, etc.).

Processor 701 may include both general and special purposemicroprocessors, and may be the sole processor or one of multipleprocessors of computer 700. Processor 701 may comprise one or morecentral processing units (CPUs), for example. Processor 701, datastorage device 702, and/or memory 703 may include, be supplemented by,or incorporated in, one or more application-specific integrated circuits(ASICs) and/or one or more field programmable gate arrays (FPGAs).

Data storage device 702 and memory 703 each comprise a tangiblenon-transitory computer readable storage medium. Data storage device702, and memory 703, may each include high-speed random access memory,such as dynamic random access memory (DRAM), static random access memory(SRAM), double data rate synchronous dynamic random access memory (DDRRAM), or other random access solid state memory devices, and may includenon-volatile memory, such as one or more magnetic disk storage devicessuch as internal hard disks and removable disks, magneto-optical diskstorage devices, optical disk storage devices, flash memory devices,semiconductor memory devices, such as erasable programmable read-onlymemory (EPROM), electrically erasable programmable read-only memory(EEPROM), compact disc read-only memory (CD-ROM), digital versatile discread-only memory (DVD-ROM) disks, or other non-volatile solid statestorage devices.

Input/output devices 705 may include peripherals, such as a printer,scanner, display screen, etc. For example, input/output devices 704 mayinclude a display device such as a cathode ray tube (CRT), plasma orliquid crystal display (LCD) monitor for displaying information to theuser, a keyboard, and a pointing device such as a mouse or a trackballby which the user can provide input to computer 700.

One skilled in the art will recognize that an implementation of anactual computer or computer system may have other structures and maycontain other components as well, and that FIG. 7 is a high levelrepresentation of some of the components of such a computer forillustrative purposes.

The foregoing Detailed Description is to be understood as being in everyrespect illustrative and exemplary, but not restrictive, and the scopeof the invention disclosed herein is not to be determined from theDetailed Description, but rather from the claims as interpretedaccording to the full breadth permitted by the patent laws. It is to beunderstood that the embodiments shown and described herein are onlyillustrative of the principles of the present invention and that variousmodifications may be implemented by those skilled in the art withoutdeparting from the scope and spirit of the invention. Those skilled inthe art could implement various other feature combinations withoutdeparting from the scope and spirit of the invention.

The invention claimed is:
 1. A method, comprising: dividing a frequencyrecovered signal into a plurality of blocks; for each respective blockof the plurality of blocks: estimating a carrier phase for a particularblock different from the respective block; determining values of thefrequency recovered signal for the respective block by removing thecarrier phase for the particular block from the respective block;estimating coefficients using the values of the frequency recoveredsignal for the respective block as a reference signal; and performing aphase recovery on the frequency recovered signal for the respectiveblock using a filter with the coefficients to generate a phase recoveredsignal.
 2. The method as recited in claim 1, further comprising:updating the coefficients using the phase recovered signal for therespective block; and performing an additional phase recovery on thefrequency recovered signal for the respective block using the filterwith the updated coefficients.
 3. The method as recited in claim 1,wherein at least one block of the plurality of blocks includes symbolsthat are also included in a neighboring block.
 4. The method as recitedin claim 1, wherein performing the phase recovery comprises: equalizingthe frequency recovered signal for the respective block using the filterwith the coefficients to generate the phase recovered signal; anddetermining values of the phase recovered signal for the respectiveblock.
 5. The method as recited in claim 1, wherein estimatingcoefficients comprises estimating the coefficients using a least squarebased algorithm.
 6. The method as recited in claim 1, further comprisinggenerating the frequency recovered signal by: compensating a signalusing a linear filter to generate a first compensated signal; conductinga polarization recovery and a residual dispersion compensation on thefirst compensated signal using an adaptive equalizer to generate asecond compensated signal; estimating a frequency offset by conducting acarrier frequency recovery on the second compensated signal; andremoving the frequency offset from the second compensated signal using aone-tap filter to generate the frequency recovered signal.
 7. The methodas recited in claim 6, wherein the filter comprises a fast-adaptivemulti-tap digital filter that has an adaptation rate that exceeds anadaptation rate of the adaptive equalizer.
 8. The method as recited inclaim 1, wherein the filter comprises a 1×1 linear filter.
 9. Anapparatus, comprising: a processor; and a memory to store computerprogram instructions, the computer program instructions when executed bythe processor, cause the processor to perform operations comprising:dividing a frequency recovered signal into a plurality of blocks; foreach respective block of the plurality of blocks: estimating a carrierphase for a particular block different from the respective block;determining values of the frequency recovered signal for the respectiveblock by removing the carrier phase for the particular block from therespective block; estimating coefficients using the values of thefrequency recovered signal for the respective block as a referencesignal; and performing a phase recovery on the frequency recoveredsignal for the respective block using a filter with the coefficients togenerate a phase recovered signal.
 10. The apparatus as recited in claim9, the operations further comprising: updating the coefficients usingthe phase recovered signal for the respective block; and performing anadditional phase recovery on the frequency recovered signal for therespective block using the filter with updated coefficients.
 11. Theapparatus as recited in claim 9, wherein at least one block of theplurality of blocks includes symbols that are also included in aneighboring block.
 12. The apparatus as recited in claim 9, whereinperforming the phase recovery comprises: equalizing the frequencyrecovered signal for the respective block using the filter with thecoefficients to generate the phase recovered signal; and determiningvalues of the phase recovered signal for the respective block.
 13. Acomputer readable medium storing computer program instructions, which,when executed by a processor, cause the processor to perform operationscomprising: dividing a frequency recovered signal into a plurality ofblocks; for each respective block of the plurality of blocks: estimatinga carrier phase for a particular block different from the respectiveblock; determining values of the frequency recovered signal for therespective block by removing the carrier phase for the particular blockfrom the respective block; estimating coefficients using the values ofthe frequency recovered signal for the respective block as a referencesignal; and performing a phase recovery on the frequency recoveredsignal for the respective block using the coefficients to generate aphase recovered signal.
 14. The computer readable medium as recited inclaim 13, the operations further comprising: updating the coefficientsusing the phase recovered signal for the respective block; andperforming an additional phase recovery on the frequency recoveredsignal for the respective block using the updated coefficients.
 15. Thecomputer readable medium as recited in claim 13, wherein performing thephase recovery comprises: equalizing the frequency recovered signal forthe respective block using the coefficients to generate the phaserecovered signal; and determining values of the phase recovered signalfor the respective block.
 16. The computer readable medium as recited inclaim 13, wherein estimating coefficients comprises estimating thecoefficients using a least square based algorithm.
 17. The computerreadable medium as recited in claim 13, the operations furthercomprising generating the frequency recovered signal by: compensating asignal using a linear filter to generate a first compensated signal;conducting a polarization recovery and a residual dispersioncompensation on the first compensated signal using an adaptive equalizerto generate a second compensated signal; estimating a frequency offsetby conducting a carrier frequency recovery on the second compensatedsignal; and removing the frequency offset from the second compensatedsignal to generate the frequency recovered signal.